Multiple channel FM stereo system

ABSTRACT

A transmitter or a receiver processes four-channel stereo frequency-modulation information. The information represents four audio signals A, B, C and D that correspond to sources respectively located at the left-front, right-front, left-rear and right-rear of a listening point. First and second sub-carrier signals ω s  and ω s2  both have frequencies substantially higher than the highest audio signal component. In one disclosed embodiment all of the different signals are combined to develop a signal having a carrier signal frequency that is modulated by double-sideband amplitude-modulated suppressed-carrier sub-carrier signals as expressed by the modulation function 
     
         M(t) = K.sub.1 (A+B+C+D) + K.sub.2 (A-D) cosω.sub.s t + K.sub.3 (B-C) 
    
      sinω s  t + K 4  [(A+D) - (B+C)] cosω s2  t 
     where K 1  to K 4  are constants and t is time. An alternative system provides for single sideband transmission and reception of the sub-carrier ω s2  and places an SCA channel at the location of the missing sideband.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of my continuation-in-part copendingapplication Ser. No. 401,926, now patent No. 3,944,747 filed Sept. 28,1973, which is a continuation-in-part of my application Ser. No.283,464, filed Aug. 24, 1972, now abandoned, which, in turn, is acontinuation-in-part of my application Ser. No. 190,008, filed Oct. 18,1971, now abandoned, all of which applications are assigned to theassignee of this application.

BACKGROUND OF THE INVENTION

The present invention relates to multiple-channel frequency-modulationstereo systems. More particularly, it pertains to methods and apparatusfor encoding and decoding multiple channel stereo signals.

Present-day broadcast FM stereo features the transmission of atwo-channel coherent stereo signal the modulation function of which maybe represented:

    M' (t)= K' (L+R)+ K" (L-R)sinω .sub.s t,              (1)

where L represents a left-side audio signal, R represents a right-sideaudio signal, ω_(s) is the frequency of a suppressed carrieramplitude-modulated sub-carrier signal, t is time, and K' and K" areconstants. A two-channel stereo receiver responds to a stereo broadcastby demodulating the sum and difference audio terms and then matrixingthose two terms in order to yield the fundamental left and right audiosignals L and R. The same receiver will respond to a monaural FMbroadcast by reproducing the same monaural audio signal in both of itsoutput channels. On the other hand, a monaural FM receiver will respondto the two channel broadcast stereo signal by deriving only the sum term(L+R) as represented in equation (1) and reproducing an audio signalthat represents the monaural program. The two-channel signal thus isfully compatible with the monaural signal so that a receiver properlydesigned for one also will receive the other. Further detaileddiscussion of the foregoing two-channel transmission system andexemplary disclosures of transmitters and receivers for use therewithwill be found in U.S. Pat. Nos. 3,257,511-Adler et al.;3,257,512-Eilers; 3,129,288-DeVries and 3,151,218-Dias et al, allassigned to the same assignee as the present application.

In the last few years, interest has been evident in tape-recordingsystems wherein a four-channel stereo signal is recorded on magnetictape. Four different audio signals are individually recorded on fourrespective different tracks along the tape. The four different audiosignals represent sources respectively located at the left-front,right-front, left-rear and right-rear of an originating point. By usingfour different pick-up and amplification systems together with fourseparate loudspeakers similarly distributed around a listening point,four-channel reproduction is obtained.

The advent of four-channel stereo recording and reproduction hasnaturally led to consideration of the desirability of transmitting andreceiving four-channel stereo signals by radio. Because two-channelstereo is now being broadcast by many FM transmitting stations,attention has been directed particularly to the possibility of utilizingbroadcast stations in that category of service for the transmission offour-channel stereo in addition to, or instead of, the transmission oftwo-channel stereo or monaural signals. To accomplish this requires thedevelopment of a different overall transmission signal in order toaccommodate the additional information components necessary to conveyfour separate channels. At the same time, it is desirable that anyfour-channel approach be fully compatible both with two-channel stereoand monaural, so that receiver obsolescence is avoided.

It is also desirable, from the standpoint of broadcast stationeconomics, that a commercial four-channel stereo system provide for anSCA (Subsidiary Communications Authorization) channel.

OBJECTS OF THE INVENTION

Accordingly, it is a general object of the present invention to providea new and improved four-channel stereo FM broadcast system which iscompatible with conventional two-channel and monaural broadcasting.

Another object of the present invention is to provide a four-channelstereo broadcast system in which the arrangement of the channels isconsistent with that of present-day four-channel stereo recordings.

A further object of the present invention is to provide a compatiblefour-channel stereo broadcasting system in which bandwidth requirementsare consistent with existing broadcast standards.

A related object of the present invention is to provide a stereobroadcasting system capable of permitting transmission and reception ofeither three-channel or four-channel stereo information.

It is yet another object to provide an improved four-channel FM stereobroadcast system which provides for an SCA channel.

Specific objects of the present invention include the provision oftransmitters and receivers operable in broadcast systems meeting thepreceding objectives.

BRIEF DESCRIPTION OF THE DRAWINGS

The features of the present invention which are believed to be novel areset forth with particularity in the appended claims. The organizationand manner of operation of the invention, together with further objectsand advantages thereof, may best be understood by reference to thefollowing description taken in conjunction with the accompanyingdrawings, in the several figures of which like reference numeralsidentify like elements, and in which:

FIG. 1 is a layout of a four-channel stereo-reproduction system;

FIG. 2 is a layout of a three-channel stereo-reproduction system;

FIG. 3 is a block diagram of a four-channel FM stereo transmitter;

FIG. 4 is a plot of switching waveforms preferably utilized in thesystem of FIG. 3;

FIG. 5 is a vector diagram illustrating phase relationships betweendifferent signal components produced by the transmitter of FIG. 3;

FIG. 6 is a spectral diagram of energy present in the signal developedby the transmitter of FIG. 3;

FIG. 7 is a block diagram of a four-channel stereo FM receiver capableof reproducing the signal information present in the signal transmittedfrom the apparatus of FIG. 3;

FIG. 8 is a block diagram of a three-channel or four-channel stereoreceiver responsive to a signal of a kind developed by a modified formof the transmitter in FIG. 3;

FIGS. 9 and 10 are alternative layouts of stereo-reproduction systems;

FIG. 11 is a partially schematic block diagram of a four-channelreceiver alternative to that of FIG. 7;

FIG. 12 is a detailed schematic diagram of a portion of the receiver ofFIG. 11;

FIG. 13 is a partially schematic block diagram of a four-channel stereotransmitter alternative to that of FIG. 3;

FIG. 14 is a partially schematic block diagram of a receiver similar tothat of FIG. 11 but particularly arranged to reproduce a three-channelor four-channel stereo signal from a three-channel stereo broadcast;

FIG. 15 is a block diagram of a four-channel stereo transmitter similarto the FIG. 3 transmitter but having specific provision for transmittinga four-channel stereo signal with an SCA (Subsidiary CommunicationsAuthorization) signal;

FIGS. 16, 17 and 17A are diagrams useful in understanding a "phasing"method of single sideband modulation or demodulation employed by theFIG. 15 transmitter;

FIG. 18 illustrates a four-channel stereo transmitter similar to theFIG. 13 transmitter but having the capability of transmitting a signalwhich includes an SCA signal;

FIG. 19 is a block diagram adumbrating a decoding system for use in afour-channel stereo receiver capable of decoding a four-channel stereosignal as may be transmitted by the FIG. 15 or FIG. 18 transmittershaving an SCA signal component;

FIGS. 20 and 21 are block diagram representations of alternativereceivers employing the "phasing" method for reproducing four-channelstereo information from a transmitted signal having an SCA channel; and

FIG. 22 depicts in block diagram form a four-channel stereo receiverembodiment which employs a bandstop filter and frequency divisionmultiplexing to achieve single sideband demodulation.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 illustrates a typical four-channel stereo listening area layout.Spaced generally around a circle surrounding a listening point 20 are aplurality of loudspeakers 21, 22, 23 and 24. Adopting nomenclaturealready somewhat standard in the tape recording industry, the four soundchannels are designated A, B, D and C, respectively. The four differentchannels then correspond to track numbers 1, 3, 4 and 2, alsorespectively. With a listener 25 located in the vicinity of point 20and, as illustrated, facing in a direction so that channel A is at hisleft-front, channels B, D and C are then at his right-front, right-rearand left-rear, respectively.

With the audio signals in the four channels having originally beenpicked up by a correspondingly-disposed plurality of four microphonesarranged in a circle around the source of sound, listener 25 findshimself encircled with sound in a manner characteristic of four-channelstereo reproduction. Utilizing the correlation of the audio signals asbetween channels C and D, listener 25 senses an apparent or phantomsource to his rear. That is, to the listener it is as if there were asource 26, as indicated by dashed lines in FIG. 1, positioned behind thelistener and consisting of the correlated components of channels C andD.

FIG. 1, then, suggests one general manner of transmitting and/orreproducing signals in but three channels in order to achieve what, tothe listener, is an effect of four-channel stereo. That is, the threeseparate channels necessary to be broadcast or recorded could be (A),(B) and (C+D). One approach to broadcasting such a combination ofsignals is to formulate a signal having a modulation function like thatof equation (1) in the introduction and wherein A becomes L and Bbecomes R. In addition, the combined audio signal (C+D) is impressedupon the sub-carrier in phase quadrature to the (L-R) term; that is,equation (1) would then include an additional term: (C+D) cosω_(s) t.

FIG. 2 is an improved loudspeakers layout for the reproduction offour-channel stereo information from a three-channel system of recordingor broadcasting. As will become apparent from a consideration of systemsyet to be discussed, the arrangement of FIG. 2 permits the use ofbroadcast systems wherein compatibility is obtained. In thisarrangement, a loudspeaker 30 to the left-front of listening point 20and listener 25 reproduces (A+C) audio signals. At the right-front oflistener 25 is a loudspeaker 31 that produces (B+D) audio signals.Finally, a loudspeaker 32, located to the rear of listener 25 reproduces(C+D) audio signals. Again utilizing only fully correlated signalcomponents, a phantom source 33, productive of audio signal D, appearsto the listener as if located adjacent to his right-rear. Similarly, thelistener "hears" an audio signal C arriving from a phantom source 34located adjacent to his left-rear. This occurs because of the arrival athis right ear of "D" signal information from both loudspeakers 31 and 32which are equally spaced around the circular arrangement from theposition of phantom source 33. Similarly, the listener's left earderives the combined "C" information present in both loudspeakers 30 and32.

Utilizing the arrangement of FIG. 2 for the reproduction of signalsrecorded on magnetic tape, it will be apparent that only three tracks ofrecorded signals are necessary on the tape itself in order to reproducea form of four-channel stereo in response to the pickup of four separatesignal components at locations correspondingly spaced around theoriginal sound source. In that case, the four signals originally pickedup would be combined into the three signals (A+C), (B+D) and (C+D)before recording. Analogously, the three signals so combined might beinterrelated into a modulation function for transmission by radio in aknown multiplex transmission technique such as one of the numerousapproaches presently in use for telemetry or by means of modification ofthe current two-channel stereo broadcast signal in the manner discussedabove with respect to FIG. 1. Particularly for the purpose of FM stereobroadcasting, however, a four-channel FM carrier modulation function ofa kind presently to be described is preferred because of compatibilitywith existing FM broadcast modes of transmission while at the same timepermitting a choice of either four-channel stereo reproduction asdepicted in FIG. 1 or three-channel stereo reproduction as depicted inFIG. 2.

To those ends, the transmitter of FIG. 3 includes four distinct audiosignal sources 41, 42, 43 and 44, respectively producing audio signalsA, C, D and B. Those signals again represent pick-up points locatedgenerally in a circle around an original sound source in positionscorresponding to those of the loudspeakers productive of the samerespective audio signals with reference to listening point 20 in FIG. 1.Alternatively, audio sources 41-44 may be corresponding pickup apparatusof a four-channel record mechanism such as a four-track tape recorder asdescribed above. In that case, of course, the information stored in thefour tracks originally is derived from a plurality of sound pickupsarranged at locations corresponding to those of FIG. 1.

Audio sources 41-44 are individually coupled to a correspondingplurality of pre-emphasis networks 45-48 in order to derive an improvedsignal-to-noise ratio as well understood in the art. That is, thelow-frequency portion of each audio signal, relative to thehigh-frequency components thereof, is attenuated. From the pre-emphasisnetworks, each of the audio signals then is individually fed to arespective one of gates 50, 51, 52 and 53. When gated on, the differentaudio signals all are coupled into an adder network 54 in which thedifferent gated audio signals are combined and then fed through alow-pass filter 55 to frequency modulate a main carrier developed by agenerator 56 which feeds a composite broadcast signal to an antenna 57.

A pilot or control signal developed by an oscillator 60 is doubled infrequency by a frequency multiplier 61 and fed through a phase adjustor62 to a matrix and pulse shaper 63. Particularly in the interest ofachieving compatibility with current broadcast standards for two-channelFM stereo broadcasting, the pilot signal from oscillator 60 is assigneda frequency of 19kHz and multiplier 61 is adapted to double thefrequency of this signal. Consequently, the frequency ω_(s) of thesignal fed to matrix and pulse shaper 63 is 38kHz. The function ofmatrix and pulse shaper 63 is to develop four different pulse trains,one for each different one of gates 50-53. The pulse trains areinterrelated in time as shown in FIG. 4 wherein the four different pulsetrains are labelled A, B, D and C in respective correspondence to theirapplication to gates 50, 51, 52 and 53. In more detail, matrix and pulseshaper 63 thus develops a total of four switching waveforms which haverespective Fourier series expansions as follows:

Channel A: 1/4 + 1 /π (√2 cosω_(s) t+ cos 2ω_(s) t+ . . .),

Channel B: 1/4 + 1/π (√2 sinω_(s) t- cos 2ω_(s) t+ . . .),

Channel D: 1/4- 1/π (√2 cosω_(s) t+ cos 2ω_(s) t+ . . .), and

Channel C: 1/4 - 1/π (√2 sinω_(s) t- cos 2ω_(s) t+ . . .).

Combining the audio signals produced in each channel by their respectiveswitching waveforms, and gathering terms, the composite signal in adder54 becomes:

    M(t)= K.sub.1 (A+B+C+D)+ K.sub.2 (A-D) cosω.sub.s t+ K.sub.3 (B-C) sinω.sub.s t+ K.sub.4 [(A+D)- (C+B)]cosω.sub.s2 t+ . . ., (2)

where, t is time and K₁, K₂, K₃ and K₄ are 1, √ 2, √2, and 1,respectively. In this case, ω_(s2) = 2ω_(s). It will thus be seen thatthe composite signal contains a main channel which is the sum of allfour audio channels, the difference between audio channels A and Dmodulated on an in-phase sub-carrier, a quadrature sub-carrier modulatedby the difference between audio channels B and C and a term in the formof a second harmonic of the basic sub-carrier and which is modulated bythe difference between the sum of audio channels A and D and the sum ofaudio channels C and B. Higher order terms of the Fourier expansion alsoare present but need not be utilized.

FIG. 5 is a phasor diagram showing the in-phase and quadraturesub-carrier components. Assigning zero degrees as the phase of a vector65, representing the (B-C) sub-carrier component, the (A-D) sub-carriercomponent necessarily appears at a position displaced by 90 degrees asindicated by a vector 66. By detecting along the 45 degree axis asindicated by a vector 67, a component may be derived which, ignoringmagnitudes, is the difference between the sum of the front channels andthe sum of the rear channels, thusly (A+B)-(C+D). On the 135 degree axisas indicated by a vector 68, the quantity (A+C)-(B+D) may be detected;this is the difference between the sum of the left-front and left-rearchannels and the sum of the right-front and right-rear channels.

For compatibility so that a conventional two-channel FM stereo receivermay properly operate upon such a signal, that receiver must be able todetect the combination of the two left-hand channels as the left siganl(L) and the combination of the two right-hand channels as the rightsignal (R). To that end, a 19kHz pilot signal K₅ S" also is transmitted,and its phase is adjusted so that its second harmonic has a phasecorresponding to the 135 degree axis of vector 68. The conventionaltwo-channel receiver derives the quantity (A+B+C+D) from the maincarrier and the quantity (A+C)-(B+D) from the sub-carrier. Its additivematrix then sums those two quantities to provide an (A+C) signal whichconstitutes the left (L) two-channel stereo signal. Analogously, itssubtractive matrix produces the difference between the two derivedquantities which is of the form (B+D) that constitutes the right (R)signal. Since all four of the audio signals are represented in themodulation component (A+B+C+D) upon the main carrier, it is readilyapparent that a conventional monophonic FM receiver will appropriatelyderive and reproduce a compatible monaural signal. Thus, from aqualitative standpoint the modulation function of equation (2) isarranged for compatibility as between monaural, two-channel andfour-channel stereo reception.

The above analysis of the manner in which a two-channel stereo receiverresponds compatibly to the four-channel modulation function of equation(2) also reveals that a particular sequence of switching between thedifferent channels must be employed. As specifically indicated in FIG.4, the sequence will be observed, by reference again to FIG. 1, to be ina clockwise direction around listening point 20. Alternatively, theswitching sequence may be in the opposite or counter-clockwise directionor in a zig-zag pattern as left-front to left-rear to right-front toright-rear. In any event, satisfaction of the compatibility requirementrequires that two adjacent time samples be left-channel samples orright-channel samples.

As indicated, a monophonic receiver responds simply to the main-carriermodulation to reproduce a signal of the form (A+B+C+D), the first termof equation (2). A two-channel stereo receiver derives and matrixes boththat main-channel modulation term and the audio information representedby vector 68 in FIG. 5. Accordingly, the arithmetical operationsperformed by a two-channel receiver may be as follows:

    (A+B+C+D)+ (A+C)- (B+D)= 2(A+C),

and

    (A+B+C+D)- (A+C)+ (B+D)= 2(B+D).                           (3)

for three-channel reception, in addition to the already describedcomponents necessary to the reception of a two-channel stereo signal, athird signal component is included which is the sum of the left-rear andright-rear channels. With reference again to equation (2), it will beseen that this is achieved by the provision, in addition to the mainchannel signal represented by the first term, of both the in-phase andquadrature subcarrier components represented by the second and thirdterms. That is, a complete three-channel stereo modulation function isrepresented by the first three terms of equation (2) and, for thepurpose of transmitting only a three-channel stereo broadcast, thefourth term of equation (2) may be omitted. This particular form ofthree-channel composite signal results in an interleaving of thespectral energy in the different signal components in a manner analogousto the manner of spectral interleaving as described in the aforesaidAdler et al patent for the case of the two-channel signal. Moreover, astandard SCA background channel also may be transmitted at 67kHz aspermitted under present commercial broadcast standards without anysignal degradation. In addition to the main channel modulation, thethree channel receiver then derives the audio signal information alongthe 45 degree and 135 degree axes as represented by vectors 67 and 68 inFIG. 5. The three reproduction signals as represented in FIG. 2 may thenbe derived by arranging the receiver to perform the followingarithmetical operations:

    (A+B+C+D)+ (A+C)- (B+D)= 2(A+C),

    (a+b+c+d)- (a+c)+ (b+d)= 2(b+d),

and

    (A+B+C+D)- (A+B)+ (C+D)= 2(C+D).                           (4)

it is then for the purpose of also permitting four-channel receptionfrom the same composite signal that a still additional quantity ofinformation is included within the modulation function of equation (2).While an equivalent component may be utilized, that added quantity ofinformation is contained in the fourth term of equation (2) whichpreferably involves a second harmonic of the second term of the sameequation. Consequently, a four-channel receiver responds to thecomposite modulation function by deriving the (B-C) and (A-D) signalcomponents as represented at vectors 65 and 66 in FIG. 5, the samemain-channel component (A+B+C+D) as before and also the second harmoniccomponent modulation (A+D) - (B+C). The finally desired four separateaudio stereo signals may then be derived in a four-channel receiver bymatrixing to perform the following arithmetical operations:

    (A+B+C+D)+ 2(A-D)+ (A+D)- (C+B)= 4A,

    (a+b+c+d)+ 2(b-c)+ (c+b)- (a+d)= 4b,

    (a+b+c+d)- 2(b-c)+ (c+b)- (a+d)= 4c,

and

    (A+B+C+D)- 2(A-D)+ (A+D)- (C+B)= 4D.

accordingly, it is now evident that the complete modulation functionrepresented by equation (2) permits appropriate reception by any ofmonaural, two-channel, three-channel or four-channel stereo receivers.At the same time, by omitting the final term, the same modulationfunction may be utilized for the purpose of compatibly broadcasting athree-channel stereo signal.

For clarity of understanding, it should be noted that, whenever capitalletters have been used to indicate audio signals in the above equations,they are but shorthand representations for the full audio function. Thatis, the A audio component, for example, would more completely berepresented by the expression A_(m) cos_(A) t. However, it is notnecessary to an understanding of the present invention to complicate therelationship additionally by actually including such a full statement ineach equation.

While any receiver might be self-generative of the sub-carrier signalsnecessary to derive the sub-carrier modulation information, it is ofcourse desirable that a pilot signal, as already mentioned, betransmitted along with the remainder of the broadcast information.Accordingly, the transmitter of FIG. 3 feeds a portion of the referencesignal developed by oscillator 60 through a phase adjustor 70, anamplitude control 71 and a switch 72 to adder 54 for inclusion in theultimate composite signal. Consistent with current two-channel stereobroadcast standards, amplitude control 71 is utilized to obtain a pilotmodulation of between 8 and 10 per cent. Phase adjuster 70, of course,is utilized to properly locate the pilot phase relative to thesub-carrier phases as represented in FIG. 5. Switch 72 is closed onlyduring the broadcasting of stereo information so as to enable the use ofreceiver circuitry selectively responsive and indicative as betweenmonaural and stereo operations.

In a similar manner, the once-doubled pilot signal from phase adjuster62 is fed through a frequency multiplier 74, a phase adjuster 75,another amplitude control 76 and another switch 77 to adder 54.Multiplier 74 again doubles the pilot signal so that a 76kHz secondpilot signal also is transmitted as part of the composite modulationfunction. Switch 77 is closed only during the transmission offour-channel stereo information so as to permit the receiver to beselectively responsive to and indicative of the receipt of four-channelsignals. Amplitude control 76 is employed to adjust the modulationpercentage of the second harmonic pilot signal as desired or required bybroadcast standards; this percentage again may be of the order of 8-10percent.

Where it also is contemplated sometimes to broadcast only three-channelstereo information, a still-different and additional pilot signal mayalso be fed to adder 54 so as to provide a receiver the means todistinguish between three-channel and four-channel broadcasting as wellas between three-channel and two-channel broadcasting. For example, thatstill additional pilot signal could be assigned a frequency of 38kHz inthe system described.

As has been shown, a key function in the transmitter of FIG. 3 is thatof operating upon the four different audio signals in order sequentiallyto gate different ones of those signals to the combining meansrepresented by adder 54 in a manner so as to produce the sub-carriersignals represented in the modulation function of equation (2). FIG. 6depicts the resultant spectral response of this process as applied toone channel only. Since the result is basically a Fourier seriesexpansion of a rectangular wave, the envelope amplitude as representedby dashed line 80 defines a (sin x)/x function. With each of thespectral components being modulated by an audio signal, which occurs inthe process of gating the different audio signals on and off, furtherspectral components are introduced as audio sidebands around eachfundamental, spectral component of the rectangular wave. With aswitching or gating signal of 38kHz as employed in FIG. 3, thefundamental spectral components occur at 38kHz, 76kHz, 114kHz and ateach successive higher harmonic of 38kHz within additional lobes ofenvelope 80 each of successively lower amplitude.

Thus, envelope 81 represents the audio sidebands of a suppressedsub-carrier 82 located at 38kHz. The correlated information within theaudio sidebands represented by envelope 81 is that of the second andthird terms of equation (2). A similar envelope 83 centered about asuppressed sub-carrier 84 at 76kHz represents the audio informationcontained in the fourth term of equation (2). At the same time, the mainchannel 85 is modulated by a function representative of the first termof equation (2) and indicated by envelope 86 in FIG. 6. Insofar as thefour-channel stereo information is concerned, all audio informationcontained in the sidebands attendant to the fundamental spectralcomponent harmonics of a frequency higher than 76kHz are redundant.Consequently, the spectral energy above envelope 83 preferably istruncated, as indicated by vertical dashed line 88, by means ofattenuation in low-pass filter 55 of FIG. 3.

As indicated in FIG. 6, both the 38kHz and 76kHz sub-carriers aredouble-sideband amplitude modulated. Further, it will be noted that theaudio modulating coefficient of each of the three sub-carrier terms inequation (2) is a difference between two fundamental audio quantities.Each such term is a mathematical representation of the amplitudemodulation of a carrier wherein the carrier itself is suppressed. Thatis, when the modulation vanishes the multiplying coefficient is zero asa result of which the carrier also vanishes. With a maximum audiomodulating frequency of 15kHz for each different signal component, itwill be observed that the minimum channel width required for the firstsub-carrier bands is 38kHz ± 15kHz, or from 23kHz to 53kHz. Similarly,the minimum channel width required for the second or harmonicsub-carrier sideband is from 61 to 91kHz. Thus, the spectrum preferablyis cut off between 91 and 99kHz.

Observation of the composite signal by use of an oscilloscope revealsthat, ignoring the small effect of the pilot sub-carriers, maximumpeak-to-peak amplitude of the composite modulation function is nogreater than the maximum of any of the four terms in equation (2) whencoefficients K₁, K₂, K₃ and K₄ are suitably chosen. Such inspectionshows that the main channel or first term has a maximum peak-to-peakamplitude that if four times that of any of the individual audio signalsalone. At the same time, it may be observed that the peak-to-peakamplitude of the composite of the main and the sub-carrier terms may beequal to or less than the maximum peak-to-peak amplitude of the mainchannel waveform. Consequently, the FM generator may be fully modulatedwith the main channel audio of the first term of equation (2) and thenalso fully modulated with a composite term comprising the main and allthree sub-carrier terms without having to reduce the modulationpercentage for any component as applied to the radiated carrier. Thisproperty, called interleaving, is directly related to the manner oftime-division multiplexing of the different audio components. It also isrelated to the fact that there is correlation existing between thedifferent modulation components. Moreover, the same interleavingproperty also is obtained when, instead of the use of actualtime-sequence gating as in FIG. 3, a more direct frequency-divisionapproach is utilized to form the broadcast composite signal.

As so far presented, it has been shown by use of a multiplexing approachthat the information contained within the modulation function ofequation (2) permits compatibility as between the several differentstereo modes in terms of just what information is received. However, forcomplete compatibility in all respects, it is also necessary tocorrelate properly the amplitude of each of the different signalcomponents transmitted or received as part of the overall modulationfunction and as represented by the different terms. That is, thereshould be compatibility in the sense that, for example, a monauralreceiver, other things being the same, will respond equally either to amonaural broadcast or to a four-channel stereo broadcast. This requiresthat the different constants K₁ to K₄ in equation (2) be appropriatelyweighted. With direct sequential multiplexing as discussed with respectto FIG. 4, for a value of K₁ of 1/4, K₂ and K₃ are 2/π and K₄ is 1/π.These differences in amplitude are illustrated by the differences inheight of the different envelopes 81, 83 and 86 in FIG. 6. Forquantitative compatibility as between the different stereo modes andmonophonic reception, it is, however, necessary that the constants inequation (2) be adjusted. To this end, the amplitude of the main channelmodulation is augmented or raised, as indicated by envelope 89, to equalthat of the amplitude of the sidebands on the second harmonicsub-carrier as represented by envelope 83. That is, K₁ is adjusted tobecome 1/π, with K₂ and K₃ still being 2/π and K₄ still being 1/π, andwith 100 percent modulation, constants K₁ -K₄ become 1, √2, √2, and 1respectively. This is accomplished in the transmitter of FIG. 3 byfeeding a portion of each of the original audio signals to an adder 90which then inserts an additional fraction of the main channel signal(A+B+C+D) into the composite modulation envelope by means of summationin adder 54.

The foregoing coefficient adjustment to achieve quantitativecompatibility may be further understood by noting that, when twoadjacent time slots are transmitted, the 38kHz components are inquadrature with each other. Because of their √2 coefficient term, theyadd vectorially to produce a unity coefficient term as required fortwo-channel compatibility. Under this same condition, the secondharmonic component vanishes. For the case, then, when only one channelis being broadcast, the main channel contribution is 1/π of the total,the 38kHz is √2/π of the total and the second harmonic component is 1/πof the total. Recognizing that full modulation in this situation is 4/π,it may also be noted that, for this condition, the transmitter is beingunder modulated. Viewing the necessary coefficient adjustment in anotherlight, reference may be made to the above equations for the fourswitching waveforms which were combined to yield equation (2). Theadjustment is that of replacing the 1/4 term by a 1/π term so as to beequal to the coefficient of the second harmonic term in those waveforms.

FIG. 7 depicts a time-multiplex receiver for reciprocally gating thedifferent ones of the signal components from the transmission signalreceived from a transmitter such as that of FIG. 3, while demodulatingthe audio signal components from the different sub-carrier signals.Thus, the transmission signal intercepted by an antenna 100 is receivedby one or more radio-frequency (RF) stages 101, converted to anintermediate frequency by a first detector 102, amplified in one or moreintermediate-frequency (IF) stages 103 and fed to a discriminator andautomatic-gain-control (AGC) network 104. In a conventional manner, theAGC portion of network 104 is utilized to develop a control signal thatis fed back to govern the gain of stages 101 and 103 and hence insurethe supply of a constant-amplitude signal to the discriminator.Similarly, an AFC system 105 compares the precise frequency of theintermediate-frequency carrier with the frequency of thelocal oscillatorsignal in detector 102 for the purpose of accurately fixing the responseof the detector with respect to the frequency of the received broadcastsignal.

When receiving a four-channel stereo broadcast, the signal available atthe output of the discriminator in network 104 is the modulationfunction of equation (2) together with at least the pilot signal K₅ S".Recalling that the transmitter of FIG. 3 develops the modulationfunction by repetitively gating the four different primary audio signalssequentially onto the total composite signal, the receiver in FIG. 7 forthe four-channel stereo signal is constructed to operate in a mannerwhich basically is the reciprocal of the manner of operation of thetransmitter. That is, the modulation function from network 104 issampled by four trains of pulses like those in FIG. 4 in order directlyto develop the four different audio output signals A, C, D and B. Therate of sampling is the same as the rate of sequencing at thetransmitter, in this case 38kHz, and it is appropriate, therefore, toutilize the second harmonic of the 19kHz pilot signal for the purpose oftiming the receiver sampling rate. The receiver of FIG. 7, then,includes a filter 107 selective of the 19kHz pilot signal and followedby shaping and pulse generating circuits 108 which develop the fourtrains of pulses that are fed to a gating system 109. In response to thetiming provided by those pulse trains, gating system 109 then acts tosample the composite modulation function fed to it from network 104 intime correspondence with the original time sequencing at thetransmitter. Ignoring for a moment a certain desired correction, theresulting four separate audio signals are fed individually to respectivede-emphasis networks 110, 111, 112 and 113 that operate upon the audiosignals reciprocally to the operation of pre-emphasis networks 45, 46,47 and 48 in the transmitter of FIG. 3, this operation, as such, beingentirely conventional. After de-emphasis, the four different audiosignals are amplified in respective amplifiers 114, 115, 116 and 117following which the audio signals are fed individually to respectivedifferent loudspeakers 118, 119, 120 and 121.

By sampling with rectangular waves of the form shown in FIG. 4, aquantitative error in reproduction is encountered as between thedifferent audio quantities. This is reciprocally analogous to the errorcompensated in the transmitter of FIG. 3 by use of adder 90. To the endof equalizing that sampling error in the receiver of FIG. 7, each of thefour sampled audio signals is individually fed to its de-emphasisnetwork through a respective one of subtractors 122, 123, 124 and 125.Each subtractor is fed a portion of the main channel audio component(A+B+C+D) derived from network 104 by an attenuator 126. As will bedescribed in more detail in connection with the alternative receiver ofFIG. 11, the modified signal from attenuator 126 is of the form(1-2/π)(A+B+C+D).

Modification of the receiver of FIG. 7 in a manner to be discussed withrespect to FIG. 8 enables obtaining three or four fundamental outputaudio signals such that an acoustic effect of four-channel reproductioncan be achieved in the manner of FIGS. 2, 9 or 10. In this case, filter107 and shaping circuits 108 serve to product trains of sampling pulsesof the form used in FIG. 7, the contributions from two adjacent timesamples effecting a 45° phase shift to effect sampling or demultiplexingalong the axes represented by vectors 67 and 68 in FIG. 5. As in thecase of FIG. 7, it still is necessary to modify each of the resultingaudio signals in order to compensate or equalize for the characteristicsof the rectangular wave de-multiplexing process and thus obtain cleanlyseparated audio components.

Accordingly, FIG. 8 includes a gating system 130 which receives thecomposite modulation function from the discriminator of network 104while at the same time receiving sampling pulses of the form depicted inFIG. 4. In a first mode of operation, the audio samples obtained are inthe form represented by equations (4) so as to yield the primarythree-channel stereo signals (A+C), (C+D) and (B+D). These lattersignals are then individually fed through respective de-emphasisnetworks 131a, 131b and 131c and respective amplifiers 132a, 132b and132c to corresponding loudspeakers 133a,133b and 133c. Again for thepurpose of equalization, a portion of the main channel component(A+B+C+D) is obtained from an attenuator 134 and deducted from thesignal in each respective output path by means of subtractors 135a, 135band 135c. The loudspeakers preferably are arranged as shown in FIG. 2.

In another mode of operation, the receiver of FIG. 8 further includes anadditional output channel composed of a subtractor 135d, a de-emphasisnetwork 131d, an amplifier 132d and a loudspeaker 133d. This operationalmode is activated by the closure of a switch 136 between gating system130 and subtractor 135d and of a switch 137 between attenuator 134 andthat same subtractor. As a result, the receiver also produces an outputaudio signal (A+B), representing the arithmetical operation:

    (A+B+C+D)+ (A+B)- (C+D)= 2(A+B).                           (4a)

Accordingly, the loudspeakers are now arranged as shown in FIG. 9, sothat the quantities (A+C), (C+D), (B+D) and (A+B) appear respectively atthe left, rear, right and front of listener 25. As indicated bydashed-line loudspeaker representations in FIG. 9, this simulatesreproduction by four sources A, B, C and D arranged as in FIG. 1. Yet,information contained in only the first three terms of the modulationfunction of equation (2) has been utilized in this mode as in the oneimmediately above. That is, only one subcarrier frequency is employed asin the case of conventional two-channel stereo systems. As compared withthe latter, however, a quadrature component has also been used.Synchronization again is with respect to the axes represented by vectors67 and 68 in FIG. 5.

FIG. 10 represents a possible alternative to the arrangement of FIG. 9.In this case, loudspeakers 138a, b, c and d are arranged around listener25 as in FIG. 1 for loudspeakers 21, 22, 23 and 24, respectively. Theoutput signals fed to each of loudspeakers 138a, b, c and d are(C+2A+B), (A+2B+D), (B+2D+C) and (A+2C+D), respectively. Of course,these particular signals may be produced by simple algebraic matrixingof the signals produced by the receiver of FIG. 8 for the reproducersystem discussed above with respect to FIG. 9. In each case, theloudspeaker in FIG. 10 reproduces a predominant audio component of theFIG. 1 type at the appropriate location together with coherentcomponents balanced from the two locations on either side of thatlocation.

FIG. 11 dipicts a modification of the receiver of FIG. 7 for the purposeof developing a full four-channel stereo program. Thus, an FM detector140 represents the discriminator portion of network 104 in FIG. 7. Thedetected composite modulation function of equation (2) is fed to each offirst and second gates 141 and 142 and product detectors 143, 144, 145and 146. A filter, multiplier and matrix network 148 serves to derivethe 19kHz pilot signal K₅ S" double that signal and, by means of phaseshifting and inverting circuits, develop demodulation signals of theforms (sinω_(s) t), (cosω_(s) t), (-sinω_(s) t) and (-cosω_(s) t). Thesedemodulation signals individually are fed respectively to productdetectors 143, 144, 145 and 146. Each of the product detectors functionsas a synchronous demodulator and together they function to demodulatethe audio information within the composite modulation function ofequation (2) so as respectively to derive the information represented inthe phasor diagram of FIG. 5 respectively at 0°, 90°, 180° and 270°. Thefour product detectors 143, 144, 145 and 146 thus yield the separatedaudio components (B-C), (A-D), -(B-C) and -(A-D), respectively.

A filter and shaper network 150 extracts the 76kHz fourth harmonic ofthe 19KHz pilot signal and develops a pair of sampling signals of theform 1±(4/π) cosω_(s2) t that are fed respectively to gates 141 and 142.Gates 141 and 142 serve as binary or on-off switches and thus operateupon the fourth term in the modulation function of equation (2). As alsoin the case of FIG. 7, the lack of complete separation inherent in theoperation of gate 141 is equalized or compensated by combining therewitha modifying signal of the form -(1-2/π)(A+B+C+D). To this end, themain-channel audio portion of the composite modulation function fromdetector 140 is fed through an amplifier 151 and summed in an adder 152with the audio component derived from the output of gate 141. Amplifier151 is adjustable in gain so that the precise level of compensatingsignal may be obtained; amplifier 151 thereby serves as a separationcontrol. Similarly, the compensating signal from the output of amplifier151 is also combined with the audio component from the output of gate142 in an adder 153. Consequently, the modified signals sampled throughgates 141 and 142 are of the form (A+D) and (B+C), respectively.

It may be helpful to explain the reasons for the nature of theparticular demodulating processes indicated for gates 141 and 142. Asmay be deduced from the aforementioned Adler et al. patent, a signalcomponent having a form such as that of the fourth term of equation (2)may be demodulated by operating upon it with a multiplifer of the form1±2 cosω_(s2) t. In this manner, the audio information modulated uponthe fourth term of equation (2) would be derived directly by operationof the gating system. In view of the inconvenience or impracticabilityof arriving at multipliers of the form just discussed, however, use ismade of a sampling square wave for the second term of equation (2) ofthe above-mentioned form:

    1±(4/π) cosω.sub.s2 t.                         (6)

A half sine wave may also be used which corresponds to the multiplier1±(π/2) cosω_(s2) t. As also explained in the aforementioned articles,use of a multiplying or sampling waveform of the kind represented byequation (6) necessitates the already mentioned modification orequalization of the derived modulation components in order ultimately toobtain clean separation between the different primary stereo audiosignals. This process of equalization may be understood by a detailedexamination of the sampling of one phase of the fourth term in equation(2). In this case, then, the multiplying wave is of the form 1+4/π(cosω_(s2) t). The multiplication of that term with the modulationfunction of equation (2) as obtained from the discriminator in network104 may be expressed:

    {(A+B+C+D)+ [(A+D)- (B+ C)] cosω.sub.s2 t}(1+(4/π) cosω.sub.s2 t).                                     (7)

The multiplication process yields:

    (A+B+C+D)+ 4/π [(A+D)- (B+C)] cos2ω .sub.s2 t+  . . . (8)

Since the needed information is contained entirely within the indicatedfirst two terms of equation (8), the useful portion of that equation maybe expressed:

    (A+B+C+D)+ 4/π[(A+D)- (B+C)] (1/2+1/2 cos2ω .sub.s2 t)= (A+B+C+D)+2/π[(A+D)- (B+C)]= 1.673 (A+D)+ 0.363 (B+C). (9)

whereas the desired result of the multiplication process represented byequation (7) would be the development of a pure audio component of theform (A+D), it is observed that, instead, the resultant includes acontribution of (B+C), representing a separation of only 13 db.

While corrective weighting might be accomplished in different ways, adirect approach is to include amplifier 151 which supplies thecorrecting modifier. That modifier is the now familiar quantity[-(1-2/π)(A+B+C+D)]. Thus, equation (9) is modified as follows:

    (A+B+C+D)+ (2/π)[(A+D)- (B+C)+ [-(1-2/π)(A+B+C+D)]= 4/π (A+D) (10)

accordingly the process begun in equation (7) as now modified yields acompletely separated form of the audio information (A+D). By in the samemanner modifying the other resultant obtained by combining the otherpolarity of the fourth term of equation (2) with the respectivemodifier, a cleanly separated signal also is obtained of the form (B+C).

With development of the signals as thus far discussed in connection withthe receiver of FIG. 11, it will be observed that it now is necessaryonly to additively matrix a total of six audio components in order toderive the ultimately desired four primary stereo signals. The audiocomponent from each of product detector 143-146 are individually fedthrough respective adjustable attenuators 156, 157, 158 and 159 andamplifiers 160, 161, 162 and 163. Amplifiers 160-163 serve to bring upthe level of the audio components they individually handle to a valueequal to that of the audio components developed by gates 141 and 142 andthen modified. Adjustable attenuators 156-159 serve as additionalseparation controls for the purpose of enabling the clean development ofeach different signal component.

Finally, an adder 166 sums the (A+D) and (A-D) to yield the A stereosignal. An adder 167 sums the (B+C) and (B-C) signal components to yieldthe B stereo signal. Similarly, the C stereo signal is obtained bysummation in an adder 168 of the (B+C) and -(B-C) audio components,while an adder 169 sums the (A+D) and -(A-D) audio components to developthe D stereo signal. As in FIG. 7, the four stereo signals are then fedthrough their respective chains of de-emphasis networks and amplifiersto the individually corresponding loudspeakers 118-121.

Preferably, the receiver of FIG. 11 also is arranged to permitreproduction from a broadcast two-channel stereo signal as well as froma broadcast monaural signal. For monaural reception, all this requiresin principle is the provision of a channel for transmitting themain-carrier audio component (A+B+C+D) in common to all four of theoutput audio signal channels. This may be accomplished directly byhaving a channel connected between detector 140 and each of de-emphasisnetworks 110-113, while including in that additional channel a gateresponsive to the absence of the 19kHz pilot signal for passing themain-channel audio component. Accordingly, gates 141 and 142 arearranged to remain open in response to the absence of the 19kHz pilotsignal, preferably with the concurrent disablement of amplifier 151, soas to pass the main-channel audio component (A+B+C+D) through each ofadders 166-169 to all of the audio output channels. At the same time,each of product detectors 143-146 are arranged to operate in a mannersuch that they are disabled in the absence of their reference signals.That is, they include a threshold so as to be operative except in theabsence of an applied reference signal.

For the reception of a two-channel stereo broadcast signal, the receiverof FIG. 11 is arranged so that a demodulated and matrixed left signal Lis fed in common to the A and C output audio channels while a similarlyderived right signal R is fed in common to the B and D output audiochannels. To that end, ganged switches 170 and 171 are closed,preferably automatically in response to the absence of the 76kHz pilotsignal but in the presence of the 19kHz pilot signal, to connect inparallel the reference signal inputs of product detectors 143 and 146and 144 and 145, respectively. This forces the thusly paired productdetectors to operate on the respective axes represented by vectors 67and 68 of FIG. 5. As in the case of monophonic reception, gates 141 and142 are held open to pass the main-channel signal (L+R). In operation,detectors 143 and 146 produce a negative difference signal -(L-R), whiledetectors 144 and 145 produce a positive difference signal (L-R). Adders166 and 168 both sum an (L+R) component derived from one of gates 141and 142 with an (L-R) component derived by detectors 144 and 145respectively. In consequence, an audio component of the form (A+C) or(L) is fed to both of output channels A and C. At the same time, adders167 and 169 each sum an (L+R) signal derived from one of gates 141 and142 with a -(L-R) signal obtained by way of product detectors 143 and146, respectively. As a result, an audio signal of the form (B+D) or (R)is fed to each of output audio channels B and D. Further analysis of thecorrelated relationship between the different audio components willreveal a variety of other combinations of the circuitry that may beutilized for the purpose of performing the necessary arithmeticaloperations to energize the different ones of the loudspeakers with theappropriate ones of the left and right signals present in thetwo-channel stereo broadcasts.

FIG. 12 sets forth a more detailed schematic diagram of an audioderivation approach that may be used in conjunction with the receiver ofFIG. 11. By way of an isolation transistor stage 180 and aphase-selective tuned circuit 181, the 38kHz second harmonic of the19kHz pilot signal, in the form sinω_(s) t, is injected at the base ofone transistor of the two in each of a pair of product detectors 182 and183. A bridged-T network 184 includes a constant current source in theform of a fixed-bias transistor in its common leg for maximumcommon-mode rejection, and a pair of driver transistors individually ineach of its respective arms with the latter individually driving thecommon emitters of each of product-detector pairs 182 and 183. Fromdetector 140 through an isolation transistor stage 185, the compositemodulation function of equation (2) is applied to the base of the drivertransistor directly coupled to product-detector 182. The base of theother driver transistor is fixed biased as are the bases of thetransistors in the product detectors not coupled directly to the 38kHzreference signal. In operation, the audio component (B-C) appears on thecollectors of one transistor of each product-detector pair connected toa lead 187, while the audio component -(B-C) appears on the remainingcollectors of the two detectors that are connected to another lead 188.The bridging resistor 189 in the T-network serves as a gain control topermit adjustment of balance between the product-detector and theadditional detectors described below.

A physically similar pair of product detectors 190 and 191 areindividually driven by respective driving transistors 192 and 193 whichare included in the respective arms of another bridged-T network 194,which also includes a fixed-bias transistor in its common leg forcommon-mode rejection. The remaining bias arrangements also are the sameas network 184, and the composite modulation function from isolationstage 185 is inserted on the base of driving transistor 192. A 38kHzreference signal of the form cosω_(s) t is delivered from aphase-determining tuned circuit 195 to the bases of one transistor ineach of the product-detectors 190 and 191. In this case, the phase ofthe applied reference signal effects operation of the product detectorpairs to derive the audio quantity (A-D) on the one collector in eachproduct detector connected to a lead 197. The other collectors areconnected to a lead 198 and derive the audio component -(A-D). Again, anadjustable resistor 199 in bridged-T network 194 serves as a gaincontrol.

In the upper level of components in FIG. 12 are gate pairs 200 and 201with the emitters and bases in each pair being connected in common. A 76kHz switching signal, again derived directly or indirectly from the19kHz pilot signal, is fed through an isolating stage 203 and aphase-selective tuned circuit 204 to the bases of each of the gate pairs200 and 201. The reference signal amplitude is sufficient to saturate,and thus cut off, gate pairs 200 and 201 so as to create a switchingfunction having a multiplier of the form 1±(4/π) cosω _(s2) t. Connectedbetween the common emitters of each gate pair 200 and 201 are the armsof a resistive T network 206 with each arm including a driver transistorand the common leg having an adjustable resitor 207 serving as aseparation control. The composite modulation function from isolatingstage 185 is fed to the base of the transistor in network 206 directlyconnected to the emitters of gate pair 200, while the base of thetransistor in the other arm is returned to a point of fixed bias. Inresponse to the 76 kHz pulsed switching signal and the compositemodulation function, an audio component (A+D) appears on the collectorof each transistor in gate pair 200, while the audio quantity (B+C)appears on the collectors of each transistor in gate pair 201. Theindividual collectors of gate pair 200 are connected respectively toleads 197 and 198, and the individual collectors of gate pair 201 areindividually connected to respective leads 187 and 188. Adding thesedifferent audio components to the several audio components individuallyapplied to the respective different leads by the product detectors aspreviously described, it will be seen that the resultant audio signalson leads 188, 187, 198 and 197 are predominantly of the respective formsC, B, D and A.

However, as discussed above in connection with the rectangular wavesampling process of gates 141 and 142 in the receiver of FIG. 11, themultiplying process in gate pairs 200 and 201 results in a need forequalization or compensation by the additional summation of a modifierof the form -(1-2/π) (A+B+C+D). This is accomplished by means ofamplifier pairs 210 and 211 each composed of two transistors and withthe bases of all four transistors being returned in common to a point offixed bias. The collectors of the four transistors are individuallyconnected to the respective different ones of leads 187, 188, 197 and198. The emitters of amplifier 210 are connected to the collector of thetransistor in one arm of T-network 206, while the emitters of the otheramplifier pair 211 are connected to the collector of the transistor inthe other arm of that network. In consequence, a weighted portion of themain channel component (A+B+C+D) of the modulating function is suppliedby amplifier pairs 210 and 211 to each of leads 187, 188, 197 and 198 sothat pure primary audio stereo signals B, C, A and D are developed onthe respective leads for delivery to the respective output audiochannels as shown in FIG. 11.

As specifically arranged in FIG. 12, direct-current energization of theoverall circuitry is obtained by connecting a source of B+ throughseparate decoupling networks to all of the collectors of the differenttransistor pairs 200, 201, 210 and 211, while the common leg of eachT-network is returned to ground. The various different bases arereturned to appropriate sources of bias potential as is one terminal ofthe transistor in each isolating stage, and the emitter of thetransistor in each isolating stage also returned to ground or a plane ofreference potential.

FIG. 13 represents a broadcast transmitter alternative of that of FIG.3. In contrast with the time multiplex approach in the design of FIG. 3,however, the basis of the arrangement in FIG. 13 is that of thesynthesis of the modulation function of equation (2) by operationseparately with respect to the four different basic terms in theequation. Thus, the four different primary audio stereo signals A, D, Cand B, picked up live or derived from a four-track recording, aredelivered by respective sources 220, 221, 222 and 223 throughcorresponding pre-emphasis networks 224, 225, 226 and 227 to particularphase splitters 228, 229, 230 and 231. Each phase splitter develops apositive and negative version of the respective signal fed to it,although in this particular arrangement the negative quantity of theaudio signal A is not used.

An adder 232 sums the +A and +D signals to yield the audio components(A+D), while another adder 233 combines the +A and -D signals to yieldthe component (A-D). Similarly, an adder 234 sums the +C signal and the+B signal to produce (B+C) which, in turn, is fed to still another adder235 for combination with the (A+D) signal to yield the main-channelaudio component (A+B+C+D). The latter is fed directly to a compositesignal adder 236. Additionally, an adder 237 combines the -C and -Baudio signals to yield an audio quantity (-B-C), and a still furtheradder 238 sums the +B and -C audio signals to deliver the audiocomponent (B-C). The (A+D) and (-B-C) components respectively fromadders 232 and 237 are combined in a final adder 239 to yield the audiocomponent (A+D)-(B+C), and the latter is fed through an adjustableattenuator 240 to a multiplier 241. Similarly, the (A-D) component fromadder 233 is fed through an adjustable attenuator 243 to a multiplier244, and the (B-C) component from 238 is fed through an adjustableattenuator 246 to a multiplier 247.

For developing the different pilot and multiplying signals, thetransmitter of FIG. 13 utilizes a chain of stages basically like that inFIG. 3. That is, an oscillator 250 serves as a stable source of a 19kHzfundamental component that is fed through a phase adjuster 251, anamplitude control 252 and a switch 253 to composite signal adder 236 inorder to insert the pilot signal K₅ S". As before, switch 253 is openedwhen only the monaural component (A+B+C+D) is transmitted so as topermit appropriate response by stereo receivers.

The fundamental reference signal from oscillator 250 also is doubled ina frequency multiplier 255 and then fed through a phase adjuster 256 todevelop a subcarrier of the form sinω_(s) t which is fed to multiplier247. By operation of the latter, the third term of equation (2), K₃(B-C) sinω_(s) t, is developd and fed to composite signal adder 236. Thereference signal at the output of phase adjuster 256 also is changed inphase by 90° in a phase shifter 258 to develop a subcarrier signal ofthe form cosω_(s) t, and this latter subcarrier signal is then fed tomultiplier 244, the action of which is to produce and feed to compositesignal adder 236 the second term of equation (2) of the form K₂ (A-D)cosω_(s) t.

The reference signal at the output of shifter 258 also is again doubledin another frequency multiplier 260 and then in part fed through a phaseshifter 262 for development of the subcarrier signal cosω_(s) t. Thelatter is then supplied to multiplier 241 which functions to produce andfeed to composite signal adder 236 the last term of equation (2) in theform K₄ [(A+D)-(B+C)] cosωs2t. An additional portion of the referencesignal from frequency multiplier 260 is fed through a phase shifter 264,an amplitude control 265 and a switch 266 to composite signal adder 236.Switch 266 is open except during the transmission of four-channel stereoinformation, so that the higher-frequency pilot signal K₆ S'" is presentonly during the broadcast of that stereo mode. Finally, the now-completecomposite modulation function of equation (2) together with the twopilot signals is fed from adder 236 to frequency modulate themain-carrier developed by a generator 268 and fed to an antenna 269.

In FIG. 14 the receiver of FIG. 11 has been modified so as to enable thedevelopment of either three or four output audio stereo signals thatsimulate four-channel reproduction in the manner discussed in connectionwith FIGS. 2 and 9. To this end, the apparatus and manner of operationof the receiver of FIG. 14 is the same as that in FIG. 11 in eachdifferent signal path through amplifiers 160-163. Departing from thearrangement of FIG. 11, the main-channel audio component (A+B+C+D) isderived directly from detector 140. The latter quantity is then summedin an adder 271 with a component (A-D) from amplifier 161 and then againin an adder 272 with the component -(B-C) from amplifier 162 to yield astereo component of the form (A+C). In an adder 273, another portion ofthe main-channel component is summed with the component -(A-D) fromamplifier 163 and that combination then is further summed in an adder274 with the component (B-C) from amplifier 160 to yield a second audiostereo signal of the form (B+D). Next, the combination at the output ofadder 273 also is summed in another adder 275 with the component -(B-C)so as to yield the third stereo audio signal of the form (C+D). Thesethree audio stereo signals of the forms (A+C), (B+D) and (C+D) may thenbe fed to the respective output channels of FIG. 8 so as to permitreproduction of the corresponding different ones of these stereo signalsby loudspeakers 133a, 133b and 133c which are arranged in the manner ofFIG. 2.

As shown, the receiver of FIG. 14 also includes another adder 277 whichreceives both the main channel component and a component -(C+D) derivedfrom the output of adder 275 by way of an inverter 278. Consequently,adder 277 develops a further audio stereo signal of the form (A+B) whichthen may be fed to the corresponding audio channel in FIG. 8 thatterminates with loudspeaker 133d, all of the loudspeakers in this casebeing arranged in the manner of FIG. 9. As in the case of the receiverof FIG. 8, different matrixing schemes may be incorporated in thereceiver of FIG. 14 so as to instead utilize the reproduction pattern ofFIG. 10.

Several different transmitters and receivers have been described toillustrate the attributes of a four-channnel FM stereo transmissionsystem which also permits a mode of three-channel reproduction thatproduces an equivalent effect. Because of that equivalence, attentionhas also been directed to transmitters and receivers based initially ona three-channel mode of broadcasting and reception. At the same time,both three-channel and four-channel systems have been disclosed whichpermit full and complete compatibility with receivers designed fordifferent stereo modes as well as those that reproduce only monophonicinformation.

While particular matrixing or signal combining arrangements have beendescribed for performing different arithmetical operations according tovarious specific techniques, an analysis of the different combinationsof the fundamental stereo signals that might be made together with thespecific audio information to be contained in the broadcast modulationfunction reveals that other and different arithmetical approaches may beutilized. In a transmitter, for example, the desired three-channelmodulation components (A+C), (B+D) and (C+D) may be obtained simply byadding to the monaural component (A+B+C+D) the respective quantities-(C+D), -(A+C) and -(A+B). As another example, an alternate four-channelstereo receiver approach also begins with the development of themonaural component (A+B+C+D). The stereo signal A is then obtained byadding to that monaural component the quantities (A-D) and -(B+C).Similarly, the stereo signal B is obtained by instead adding thecomponents (B-C) and -(A+D), while the stereo signal C is produced byadding -(B-C) and -(A+D). Finally, the stereo signal D is developed bysumming the monaural signal with both the components -(A-D) and -(B+C).

In any event, the overall system concepts, as well as the specificreceiver and transmitter approaches, of the present invention not onlyprovide four-channel and three-channel stereo systems that arecompatible with conventional two-channel stereophonic and one-channelmonaural broadcasting, but the arrangements are consistent with the useof present-day four-channel stereo tape recordings as well as with thebandwidth and other requirements of existing commercial broadcaststandards. Noting also that an existing two-channel stereo receiver willproduce a portion of the information contained in a three-channel orfour-channel transmission, it is apparent that an adapting unit may becoupled into such a two-channel receiver in order to expand its functionto include the reproduction of additional channels. The addition ofstill further adapting components similarly would enable themodification of a monophonic receiver for the purpose of reproducingstereophonic broadcasts of any of the different stereo modes.Analogously, present-day monophonic or two-channel stereo broadcasttransmitters, by application of the principles of the present invention,may be modified so as additionally to be capable of broadcasting thefurther stereo modes herein presented.

There are described above a number of four-channel stereo transmitterand receiver embodiments which do not specifically provide for an SCA(Subsidiary Communications Authorization) channel. It is an object ofthis invention to provide systems and methods for communicating fourfull-bandwidth independent channels which provide spectrum space for anSCA channel.

FIG. 15 illustrates a four-channel stereo signal transmitting systemwhich is similar to the above-described FIG. 3 transmitter, but whichprovides for the inclusion of an SCA channel. Receall that the FIG. 3transmitter is described above as generating a modulation function asdefined by equation (2) above in which the second sub-carrier ω_(s2),representing a second harmonic of the first sub-carrier ω_(s) and afourth harmonic of the pilot signal, is suppressed-carrier, doublesideband amplitude-modulated by the function K₄ [(A+D)- (C+B)]. In thedescribed FIG. 3 transmitter, the described second harmonic sub-carrieris assigned a frequency (1/2π)ω_(s) 2 of 76kHz. FCC regulations allowprovision in two-channel FM stereo communication systems for an SCAchannel between 53 and 75kHz.

In accordance with one aspect of this invention a four-channel stereotransmitter is provided in which the lower sideband of the secondharmonic sub-carrier ω_(s) 2 is suppressed or removed to provide forselective transmission and reception of an SCA channel which is locatedin frequency in the position of the missing lower sideband of the secondharmonic sub-carrier. A number of systems and methods are contemplatedfor achieving the described single sideband modulation of the 76kHzsub-carrier without significantly degrading the performance of the basicfour-channel stereo communication systems, as described above.

A preferred method for achieving the desired single sideband modulation,herein termed the "phasing" method, is described generally in U.S. Pat.No. 1,666,206, and in the following publications:

1. Design of RC Wide-Band 90-Degree Phase-Difference Network; by DonaldK. Weaver, Jr., Proceedings of the IRE, p. 671-676, April, 1954;

2. Normalized Design of 90° Phase-Difference Networks; by S. Bedrosian,IRE Transactions on Circuit Theory, June, 1960, pp. 128-136; and

3. Broad-Band Passive 90° RC Hybrid. . .; by A. Rogers, IEEE ProceedingsLetters, November, 1971, pages 1617-1618.

As background for a discussion of the FIG. 15 transmission system, andother subsequently described embodiments of this invention, this phasingmethod of single sideband modulation will be described briefly and ingeneral terms, with reference to FIGS. 16 and 17. FIG. 16 shows a systemin which a simple sinusoidal input signal Acosωt is supplied to parallelcircuit branches, one of which contains a first phasing network 300, tobe described below, and a balanced modulator 302. The second circuitbranch includes a second phasing network 304 and a second balancedmodulator 306. The two circuit branches are rejoined in an adder 308.

The first and second phasing networks 300, 304 apply to the signals inthe respectve branches predetermined phase-versus-frequencycharacteristics. The said characteristics are such that a 90° phasedifference is established between the signals in the two circuitbranches, assuming phasing means 300 has least phase shift. In thepresent specification, phasing means introducing the relatively lesserphase shift are termed "P"-type; the companion phasing means is termed"N"-type. In a practical system, the phasing networks 300, 304 aredesigned to produce a 90° phase difference between the signals in thetwo branches over a selected band of frequencies. In the presentapplication the frequency band of interest is the band of audiofrequencies between about 50-15,000Hz.

FIG. 17 illustrates hypothetical phase-versus-frequency characteristics310, 312 which might be produced by the P-type and N-type phasingnetworks 300, 304, respectively. In FIG. 17 the frequency band ofinterest is ω₂ -ω₁. The balanced modulators 302, 306 are supplied withreference signals cosω_(c) t and sinω_(c) t, respectively. As shown bythe annotations on FIG. 16, upon adding the outputs of the modulators302, 306 a selected one of the sidebands of the carrier ω_(c), depictedas being the lower sideband, is cancelled, leaving only the uppersideband.

This upper sideband signal can be represented as Acosω_(c) t- Asinω_(c)t where A is an audio signal, not necessarily sinusoidal, and Arepresents the Hilbert transform of A, which means that every frequencycomponent of A is shifted 90° in phase relative to every respectivecomponent in A. For example, for A= cosωt, A= sinωt.

Referring to the phasing networks 300, 304, where network 304 (N-type)produces 90° more phase shift then network 300, it can be said thatnetworks 300, 304 have outputs A and -A, respectively.

As noted above, the FIG. 15 transmitter represents the above-describedFIG. 3 transmitter modified to provide for an SCA channel. Referencenumerals in FIG. 15 which are like those in FIG. 3 identify systemcomponents having similar function. In order to modify the FIG. 3transmitter for single sideband modulation of the second harmonic(76kHz) sub-carrier, there are inserted in the four input audio channelsfollowing the pre-emphasis networks, 45, 46, 47 and 48 like phasingnetworks 314, 316, 318 and 320; the phasing networks may be constructedas described in the above-noted prior art references. Phasing networks314, 316, 318, 320 introduce a like predetermined phase-versus-frequencycharacteristic, herein shown as being of the P-type, over apredetermined band of audio frequencies in each of the four audio inputsignals.

As described in detail above in connection with the FIG. 3 transmitter,the gates 50, 51, 52 and 53 time-multiplex the four audio input signalsto develop a modulation function which includes a 76kHz sub-carriermodulated by a function proportional to [(A+D)-(C+B)]. In the FIG. 15system the complete modulation function, including the described 76kHzsub-carrier component, is subjected to the said phase-versus-frequencycharacteristic of the like phasing networks 314, 316, 318 and 320.

To achieve cancellation of the lower sideband of the 76kHz sub-carrierso as to provide spectrum space for an SCA channel centered at about67kHz, there is provided a matrix 322 receiving as inputs the fourpre-emphasized audio input signals. The matrix 322 is so constructed asto produce an auxiliary four-element difference signal (A+D)-(B+C).

The described auxiliary difference signal produced by the matrix 322 isfed to a phasing network 324 which impresses on the difference signal asecond predetermined phase-versus-frequency characteristic, here shownas being of the "N"-type. The phase-shifted auxiliary difference signaldeveloped by the phasing network 324 is fed to a balanced modulator 326receiving a 76kHz reference signal, -sinω_(s) t, from multiplier 74which has been phase shifted by 90° in a phase shifter 328 and adjustedin phase by means of a phase adjustor 330. The output signal from thebalanced modulator 326 can be varied in amplitude by means of amplitudecontroller 327.

The phase shifted output signal may be described: -[(A+D)-(B+C)]sin2ω_(s) t. The four-element difference signal component developed bythe gates 50, 51, 52 and 53 may be described: [(A+D)-(B+C)] cos2ω_(s) t.

The adder 54' performs generally the function of the adder 54 in theFIG. 3 system, but is modified to receive the described auxiliaryfour-element difference signal. The four-element difference signals arecombined in the adder 54' to effect cancellation of the lower sidebandof the second sub-carrier, producing a resultant second harmonicsub-carrier modulation component:

    [(A+D)-(B+C)] cos2ω.sub.s t- [(A+D)- (B+C)] sin2ω.sub.s t.

An SCA signal generator 332 is illustrated as being coupled to the adder54' through a switch 334 representing means for introducing at theoption of the broadcaster an SCA signal to accompany the basicfour-channel stereo modulation function.

The FIG. 15 system can be considered as a time-division multiplexingsystem in which the four audio input signals are sampled in sequenceaccording to a prescribed four-phase switching function. The inventionis applicable also to signal encoding systems of the frequency-multiplextype wherein the signal components to be combined in a compositemodulation function are generated in a frequency domain and subsequentlycombined.

The above-described FIG. 13 transmitter may be characterized as afrequency-multiplex system. FIG. 18 illustrates the FIG. 13 transmittermodified according to this invention to incorporate an SCA channel. TheFIG. 18 transmitter when compared to the FIG. 13 transmitter featuresmodifications and additions similar to those made to the FIG. 3transmitter to obtain the above-described FIG. 15 system. Primedreference numerals in FIG. 18 corresponding to those used in FIG. 15indentify like modifications and additions. The structure and operationof the FIG. 18 embodiment are deemed to be self-evident from aninspection of FIG. 18 taken in connection with the above description ofthe FIG. 13 system and the modified FIG. 15 version of the FIG. 3system.

Whereas the FIG. 15 transmitter may be characterized as a purelytime-division multiplex system, and whereas the FIG. 18 transmitter maybe characterized as a purely frequency-division multiplex system, thisinvention comtemplates other embodiments which might be characterized ashybrids, having both time-division and frequency-division multiplexcomponents. Elaborating, the main channel and two 38 kHz sub-carriermodulation components in quadrature may be generated in such a hybridembodiment by the use of a 38 kHz two-phase gating function; the 76 kHzsub-carrier component may be generated in the frequency domain by theuse of a single sideband modulation system.

It is also contemplated that other purely time-division multiplexingsystems may be employed-- for example, systems in which two two-phasegating functions, one operating at a 38 kHz rate and the other at a 76kHz rate, generate separate modulation components which are subsequentlycombined.

An important aspect of this invention is to provide method and apparatusfor controlling the phase of second harmonic (76 kHz) sub-carriers orsub-carriers relative to the 38 kHz sub-carriers in such a manner thatthe maximum possible peak voltage of the composite four-channel stereosignal is minimized when both 38 kHz and 76 kHz components are presentsimultaneously.

In the FIG. 3 encoder the four-channel stereo signal is generated bytime-division multiplexing. As a result the four input audio signals arecaused to be time-interleaved. This means that the maximum amplitude ofthe four-channel multiplexed signal is not larger than the maximumamplitude of the arithmatical sum of the four constituting signals. Evenfor a band-limited composite stereo signal this is still very nearlytrue.

The expression for the double sideband, amplitude-modulated secondharmonic sub-carrier (76 kHz), as developed by the FIG. 3 system, is[(A+D)-(B+C) cos2ω_(s) t.

When only the upper sideband of the 76 kHz sub-carrier is present, as isthe case in the FIGS. 15 and 18 systems, this desirable interleavingproperty is only partially achieved. The maximum instantaneous amplitudeof the four-channel signal in these latter systems can exceed theindividual signal maxima. The excess was experimentally found to beapproximately 3 Db.

In accordance with one aspect of this invention it has been discoveredthat by establishing a predetermined phase relationship between the 38kHz sub-carriers in quadrature and the 76 kHz sub-carriers, the maximumpossible (worst possible situation) peak voltage of the compositefour-channel stereo signal having 38 kHz and 76 kHz components presentsimultaneously can be minimized.

The four-channel stereo baseband signal except for pilot and SCAcomponents can be represented by:

    f(t)- (A+B+C+D)+ √2 (A-D) cosω.sub.s t+ √2(B-c) sinω.sub.s t + [(A+D)-(B+C)] cos2ω.sub.s t - [(A+D)-(B+C)] sin2ω.sub.s t.                                      (11)

In this expression the upper sideband of the second harmonic sub-carrieris represented by:

    +[(A+D)-(B+C)] cos2ω.sub.s t - [(A+D)-(B+C)] sin2ω.sub.s t. (12)

In expression (12) the in-phase diagonal difference signal [(A+D)-(B+C)]modulates the cosine wave and the Hilbert transform thereof modulatesthe sine wave.

It was found experimentally that if the maximum amplitude of the secondharmonic sub-carrier, as expressed in (12), is reduced by approximately30%, the interleaving is substantially as good as for a compositefour-channel stereo signal that includes a double sideband secondharmonic sub-carrier, as developed for example by the system of FIG. 3.

The second harmonic sub-carrier upper sideband component is preferablereduced in amplitude to 0.7. Expression (12) may then be rewritten as:

    0.7 {[(A+D)-(B+C)] cos2ω.sub.s t- [(A+D)-(B+C)] sin2ω.sub.s t}. (14)

The full expression (11) then becomes:

    f(t)= A+B+C+D+√2(A-D) cosω.sub.s t +√ 2(B-C) sinω.sub.s t + 0.7{[(A+D)-(B+C)] cos2ω.sub.s t - [(A+D)-(B+C)] sin2ω.sub.s t}.                                     (15)

There has been described above a number of encoding systems and methodsfor developing a composite four-channel stereo signal in which thesecond harmonic sub-carrier is single sideband modulated to remove thelower sideband thereof, and which includes provision for an SCA channelat the location of the missing lower sideband. This invention alsocontemplates a number of systems and methods for decoding a four-channelcomposite stereo signal having an SCA channel, as described. Thepreferred systems and methods for decoding the composite stereo signalexploit the general principles of the phasing method of single sidebandmodulation. Reference will be made to FIG. 17A in connection with abrief description of the phasing method as applied to demodulation,which is essentially the inverse process of modulation.

The FIG. 17A diagram is shown as including balanced modulators 309, 311,analogous to modulators 302, 306 in the FIG. 16 diagram, P-type andN-type phasing networks 313, 315, analogous to the phasing networks 300,304 in FIG. 16, and an adder 317. As shown by the annotations of theFIG. 17A diagram it can be seen that the audio signal due to the uppersideband is present at the output, but the audio signal due to the lowersideband is cancelled in the adder 317. To simplify the annotations, atthe output of the networks 313, 315 the functions of reference carrier2ω_(c) have been omitted. It is noted that the signals in the lowersideband region need not have any particular relationship to the uppersideband signals; any arbitrary signal, or even a noise component, atthe input occupying the spectrum region of the lower sideband will becancelled at the output.

FIG. 19 illustrates in general block diagram form a demultiplexer 350receiving a composite four-channel stereo signal as may be developed bya suitable FM demodulator (not shown). In connection with the discussionof the FIG. 19 decoder, and the subsequent discussions of anotherdecoder embodiment shown in FIG. 20, assume the four-channel signalreceived by the decoder has a preferred form according with the aboveexpression (15), namely:

    A+B+C+D+√2(A-D)cosω.sub.s t +√2(B-C) sinω.sub.s t+ k[(A-B-C+D) cos2ω.sub.s t- (A-B-C+D) sin2ω.sub.s t] + Y', (15a)

where Y' is an SCA signal located in the region of the lower sideband ofthe second harmonic sub-carrier.

Let Y' = Y_(m) sin y't

Y' = Y_(m) sin (2ω_(s) -y)t,

where y is the argument of the frequency-modulated SCA signal translateddown into the audio band.

The demultiplexer 350 is shown as implementing an important aspect ofthis invention, developing outputs at five output leads 352, 354, 356,358 and 360, four of which are passed through phasing networks 362, 364,368 illustrated as being of the P-type, and the fifth of which is passedthrough a phasing network 370, illustrated as being of the N-type. Theoutputs from the demultiplexer 350 are recombined in a combiner 372. Thedemultiplexer 350 may be of a type capable of demultiplexing in eitherthe time or frequency domain, and may, in certain embodiments whereappropriate, prepare the developed outputs, as by pre-matrixing, forrecombination in the combiner 372.

The combiner 372 comprises a matrixing system for separating theinformation carried by the five inputs thereto so as to develop fourdiscrete audio output signals, A, B, C, and D and to eliminate the SCAsignal. The combiner 372 may include in certain embodiments means foradjusting the relative amplitudes of various signals as will become moreclear from the following description of specific decoder embodiments.

The FIG. 19 diagram illustrates a novel employment of four P-typephasing networks 362, 364, 366 and 368 and one N-type phasing network370 to achieve cancellation of the lower sideband of the 76 kHzsub-carrier. As will become evident from a description below of specificdecoder embodiments employing the phasing method of demodulation, theuse of four P-type phasing networks and one N-type phasing network issignificant for a number of reasons. First, a minimal number of phasingnetworks is employed. Secondly, the phasing networks used in thegreatest number (the P-type networks) may be of less complexconstruction. The more complex, higher order N-type network need only beused in one location. It may be useful to note that four P-type networksare used (rather than a single P-type network operating on thedemodulated 76 kHz sub-carrier components) for the reason that theP-type phase shift introduced in the demodulated 76 kHz component of thebaseband signal must be preserved for the main channel and demodulated38 kHz signals in order not to distort the crucial phase relationshipsnecessary for proper matrixing of these components in order to achieveseparation between the four audio output signals.

FIG. 20 illustrates a four-channel demultiplexing system exploiting thephasing method of single sideband demodulation and operation in thefrequency domain. Assume again a received signal as described byexpression (15a). In the illustrated FIG. 20 system demodulationcomponents are taken off the 38 kHz sub-carriers, by means of a pair ofbalanced demodulators 374, 376 which receive the composite four-channelstereo signal after modification by an amplitude adjustor 380.

A reference signal developed at 382 for injection into the balanceddemodulator 374 is derived by the use of the 19 kHz pilot signalextracted from the composite stereo signal. The reference signal ispreferably caused to have a phase which is effective to detect the (A-D)modulation component of the 38 kHz sub-carrier.

A reference signal developed at 384, similarly derived from the 19 kHzpilot, is injected into the balanced demodulator 376, but is caused tohave a phase effective to detect the (B-C) modulation component of the38 kHz sub-carrier.

The outputs from the balanced demodulators 374 and 376 are fed to phasesplitters 386 and 388, respectively. The amplitude attenuator 380 isadjusted such that at the outputs of the phase splitter 386 there isdeveloped signals +2(A-D) and -2(A-D). Similarly, the outputs of phasesplitter 388 are +2(B-C) and -2(B-C).

The 76 kHz in-phase modulation component is taken from the 76 kHzsub-carrier by means of a balanced demodulator 390 receiving at oneinput the composite stereo signal through an amplitude attenuator 392and at another input 394 a 76 kHz reference carrier.

Balanced modulator 390 receives a reference carrier cos2ω_(s) t, thusdemodulating the in-phase second harmonic term (A-B-C+D) and neglectingthe quadrature second harmonic term (A-B-C+D). In the same process theSCA signal Y', if present, is also demodulated.

The output of balanced modulator 390 due to the 1/k Y', (1/k)Y' cos2ω_(s) t=(1/k)Y_(m) sin (2ω_(s) -y)t cos 2ω_(s) t=(1/2k)Y_(hd) m sin(4ω_(s) -y)t-(1/2k)Y_(m) sin yt.

The second term represents the spectrum in the audio range and may berepresented by:

(1/2k)Y = (1/2k)Y_(m) sin yt, where Y is defined as Y_(m) sin yt.

The Hilbert transform of Y can be represented by Y= -Y_(m) cos yt.

The four-channel stereo signal output of demodulator 390 can berepresented by:

    LF {(1/k)xk (A-B-C+D) cos2ω.sub.s tx cos2ω.sub.s t }= 1/2(A-B-C+D)

[ lf{ a} means the low frequency (audio) component of A].

The output of the demodulator 414 may be described as:

    (1/k)LF{ [+ k (A-B-C+D) sin2ω.sub.s t+ Y' [ sin 2ω.sub.s t}= 1/2(A-B-C+D)+(1/2k)LF{2Y.sub.m sin (2ω.sub.s -y)t sin2ω.sub.s t}= 1/2(A-B-C+D)- (1/2k)Y

the outputs from the 38kHz balanced demodulators 374, 376 and the 76 kHzdemodulator 390, as well as a main channel signal derived from thecomposite stereo signal are applied to summing networks 398, 400, 402and 404. At the output of summing network 398 there is developed thesignal 31/2A+ (1/2k)Y+ 1/2 B+ 1/2C- 1/2D. At the output of summingnetwork 400 there is developed a signal 31/2 B- (1/2k)Y+ 1/2A- 1/2C+1/2D. At the output of summing network 402 there appears the signal 31/2C- (1/2k)Y+ 1/2A- 1/2B+ 1/2D. At the output of summing network 404 thereis developed a signal 31/2 D+ (1/2k)Y- 1/2A+ 1/2B+ 1/2C. The outputsignals from the summing networks 390, 400, 402 and 404 are supplied tolike P-type phase shifting networks 406, 408, 410 and 412, respectively.The P-type phase shift networks cause the signals operated on to have apredetermined phase-versus-frequency characteristic over the audio band(approximately 50 Hz-15 kHz).

In order to effect cancellation of the information in the location ofthe lower sideband of the 76 kHz sub-carrier, i.e., the SCA signal, aquadrature demodulation component on the 76 kHz sub-carrier is derivedby means of a second balanced demodulator 414 receiving at input 416 a76 kHz reference carrier, sin2ω_(s) t, developed from the transmittedpilot signal, having a phase which is 90° displaced from the referencesignal 394, i.e., cos2ω_(s) t, injected into the demodulator 390. Theoutput of the demodulator 414 may be described as the Hilbert transformof the signal derived from the modulator 390, that is, 1/2 (A-B-C+ D)-(1/2 k)Y.

The output from the balanced demodulator 414 is fed to an N-typeshifting network 418 to develop a signal 1/2 (A-B-C+ D)- (1/2 k)Y. Thissignal developed by the 76 kHz demodulator 414 is combined with theoutput signals from the P-type phase shifting networks 406, 408, 410,412 by means of adders 420, 422, 424 and 426 and a signal inverter 428.

At the output of adders 420, 422, 424, 426 there is developed fourdiscrete audio signals which are supplied to deemphasizing means 430,432, 434 and 436 to produce four audio output signals corresponding tothe four audio input signals received by the encoding apparatus at thesignal transmitter.

FIG. 21 illustrates a system which also exploits the described phasingmethod of single sideband demodulation but which utilizes time-divisiondemultiplexing rather than frequency-division demultiplexing. In theFIG. 21 system the composite stereo signal is applied to a time-divisiondemultiplexer 438 in which the incoming signal is sampled at a 76 kHzrate. This rate is established with the aid of two 38 kHz signals whichare in quadrature with each other and which are derived from the 19 kHzpilot signal. The demultiplexer 438 is controlled by a switchingcontroller 440 driven by a 38 kHz timing signal applied at 442 which isderived from the 19 kHz pilot. The switching function developed by thecontroller 440 may be described as follows: ##EQU1##

The input signal can be described as:

    f(t)= A+B+C+D+ 2(A-d) cos ω.sub.s t+ 2(B-C) sinω.sub.s t + 1/2(A-B-C+D) cos2ω.sub.s t - 1/2(A-B-C+D) sin2ω.sub.s t+ Y'.

the presence of the coefficient 1/2 in the second harmonic sub-carrierterms will be explained below. Note also that the time-multiplex relatedform of the upper sideband of the second harmonic sub-carrier, asdescribed by expression (11) is used, rather than the better interleavedform described by expression (15).

The four sampled output signals v_(i) (i= 1, 2, 3, 4) can be representedby: ##EQU2##

It can be seen that in the signals v_(i) (i=1, 2, 3, 4), A, B, D and C,respectively are represented predominantly, but not exclusively. Toeliminate Y according to the phasing method, a portion of the output ofa 76kHz quadrature demodulator is mixed with the demultiplexer output.In this process of eliminating Y, however, portions of A, B, C and D areadded. The signals v_(i) thus have to be preconditioned so that the likeunwanted terms are present in the signals but with signs opposite tothose in the 76kHz quadrature demodulated signal. This preconditioningis performed in a first matrix network 443 developing a signal v_(m)which is fed to adders 444, 445, 446, 447.

In the illustrated embodiment,

    v.sub.m =(1/2π)-1/4)(A+B+C+D).

let the outputs of the adders 444-447 be represented by

    v.sub.ii =v.sub.i +v.sub.m  (i=1, 2, 3, 4).

Then ##EQU3##

The signals v_(ii) are supplied to like phasing means 448, 449, 450 and451 of the P-type. It will now be seen that the subsequent addition of aportion of the quadrature demodulator output signal will preciselyeliminate the Y as well as the undesired stereo signals.

A 76kHz quadrature demodulator 452 is driven by f(t) and by a carriersignal -sin 2ω_(s) t. Its output, 1/4(A-B-C+D)- 1/2Y, is attenuated inamplitude controller 453 and passed through N-type phasing means 455.The output of the phasing means 455 is represented by:

    v.sub.n = (1/2π)1/2 (A-B-C+D)- (1/2π)Y.

an inverter 457 makes minus v_(n) also available. In adders 459, 461,463 and 465 the following signals are generated:

    v.sub.5 = v.sub.11 + v.sub. n =(2/π)A

    v.sub.6 = v.sub.22 - v.sub. n =(2/π)B

    v.sub.7 = v.sub.33 + v.sub. n =(2/π)D

    v.sub.8 = v.sub.44 - v.sub. n =(2/π)C,

respectively.

A careful analysis of the signal processing in FIG. 21 will show thatany multiplier other than 0.5 in the second harmonic sub-carrier of theinput signal will not achieve both elimination of the SCA signal Y andtotal separation of the A, B, C and D audio signals.

Thus it is shown that for certain forms of the four-channel stereobaseband signal including SCA, the time demultiplexing form of decodingsuch a four-channel stereo signal is feasible.

As noted above, whereas the described phasing method of single sidebandmodulation and/or demodulation is preferred, other methods for achievingsingle sideband modulation and/or demodulation are contemplated. FIG. 22depicts a system for decoding a composite four-channel stereo signalwhich includes an SCA channel in the location of the lower sideband ofthe 76kHz sub-carrier, and which employs a bandstop filter to remove theSCA channel. FIG. 22 illustrates a decoding system which is similar inmany respects to the FIG. 20 decoding system, but which employs abandstop filter 454 to remove the SCA signal, rather than employing Nand P phasing networks to achieve cancellation of the said lowersideband. As in the FIG. 20 system, the FIG. 22 system employs a pair ofbalanced demodulators 456 and 458 receiving the composite stereo signalafter appropriate amplitude adjustment means 460 and subsequentlydemodulating the first harmonic sub-carrier. The outputs from thedemodulators 456, 458 are supplied to phase splitters 462 and 464 tomake available positive and negative versions of the output signals fromeach of the demodulators 456 and 458.

In order to demodulate the 76 kHz sub-carrier, a balanced demodulator466 is provided which receives as an input the composite stereo signalafter appropriate amplitude adjustment in an amplifier 468. The balanceddemodulator 466 corresponds to the demodulator 414 in the FIG. 20system. The output from the demodulator 468 is supplied to a phasesplitter 470. As in the FIG. 20 system the outputs from the phasesplitters 462, 464 and 470, along with a main channel component derivedfrom the composite stereo signal are fed to adders 472, 474, 476 and478. At the output of the adders are developed four discrete audiooutput signals A, B, C and D. In order to effect complete separation ofthe four audio output signals, the attenuator 460 is adjusted such thatthe outputs from the balanced demodulators 456, 458 represent thesignals 2 (A-D) and 2 (B-C), respectively. The amplifier 468 is adjustedsuch that the output from the balanced demodulator 466 yields a signal1/2 (A+D-C-B).

It is well known that bandstop filters cause phase shifts of thenon-attenuated signal components adjacent to the stop band. In thepresent application the phase-distorted signals are mainly in the uppersideband of the first harmonic sub-carrier and the upper sideband of thesecond harmonic sub-carrier. These phase shifts cause, in general, afrequency dependent lack of separation in the four decoder audiooutputs. This separation can be improved by adjusting the phase of thecarriers 480, 482 and 484 injected into the demodulators 456, 458, 466,by means of phase adjustors 486, 488 and 490, respectively. In thismanner acceptable amounts of separation of A, B, C and D can beobtained.

There have been discussed above a number of embodiments of four-channelstereo encoding and decoding systems which develop the favoredfour-channel stereo multiplexed signal having an upper sideband only onthe 76 kHz sub-carrier. By way of summarization and in order to moreparticularly point out the features and unique characteristics ofcertain aspects of this invention, there are stated below specificationsfor the preferred form of the composite four-channel stereo signal. Byway of introduction, it is useful to restate the composite stereo signalform as expressed in (15) but expanded to include the SCA and pilotsignals:

    f(t)= M+S sin ((ω.sub.s /2)t+3π/8)+K.sub.1 √2(A-D) cosω.sub.s t+K.sub.1 √2(B-C) sinω.sub.s t- Tcos2ω.sub.s t+U cos2ω.sub.s t- U sin2ω.sub.s t+ V cosΩt,                                              (16)

where the four audio input signals are:

A= left-front, C= left-back, B= right-front, and D= right-back;

M=K₁ (A+B+C+D)=K₁ (L+R);

s=0.1, the amplitude of the 19 kHz pilot sub-carrier;

T=0.05, the amplitude of the second pilot sub-carrier at 76 kHz;

U=K₂ (A+D-B-C), the diagonal difference signal;

U=K₂ (A+D-B-C), the Hilbert transform of U;

v=0.1, amplitude of the SCA FM sub-carrier at nominal center frequencyof 67kHz;

K₁ =0.75; K₂ = 3(with V=0, K₁ =0.85, K₂ =0.85K.sub. 3); and K₃ =0.7(approximately).

The baseband signal, f(t) has a maximum value of unity at any instant.

In order to better illustrate the compatibility of the composite signalexpression (16 ) with the expression for two-channel stereo (1 ), thefollowing transformation may be applied to the expression (16):

    ω.sub.s t=ω.sub.t t- 3/4 π.

The transformation constitutes merely a shift in the time axis and thusω_(t) t can be replaced again by ω_(s) t. This leads to:

    f(t)=M+S sin (ω.sub.s /2)t+ P sinω.sub.s t-Q cosω.sub.s t+ T sin2ω.sub.s t-U sin2ω.sub.s t-U cos2ω.sub.s t= V cos Ωt,                                             (17)

where

P=K₁ (A+D-B-C)=K₁ (Left minus Right), and

Q=K₁ (A=B-D-C)=K₁ (Front minus Back).

A comparison of expression (17) with expression (1) for two-channelstereo communication clearly reveals the compatibility of thefour-channel systems of this invention with present two-channelstereophony.

There follows an elaborated description of the preferred form of thecomposite four-channel stereo signal according to this invention.

The main channel component consists of the sum (A+C+B+D) of theleft-front, left-back, right-front, and right-back four channel inputsignals, respectively. The main channel frequency modulates the maincarrier 85% (excluding the SCA sub-carrier).

The pilot sub-carrier at 19 kHz frequency modulates the main carrier 10%.

The first 38 kHz sub-carrier, sinω_(s) t, is the second harmonic of the19 kHz pilot sub-carrier and crosses the time axis with a positive slope(increasing main carrier frequency) simultaneously with each crossing ofthe time axis by the 19 kHz pilot sub-carrier. The first 38 kHzsub-carrier and its side-bands signal is the first 38 kHz sub-carrierdouble sideband, suppressed carrier, amplitude modulated by afour-channel input signal, [ (A+C)- (B+D)] , which corresponds to atwo-channel, left minus right (L-R) input signal. The first 38 kHzsub-carrier and its sidebands signal frequency modulates the maincarrier 85% (excluding the SCA sub-carrier).

The second 38 kHz sub-carrier, cosω_(s) t is the second harmonic of the19 kHz pilot sub-carrier and is in quadrature with the first 38 kHzsub-carrier. The second 38 kHz sub-carrier causes an upward peakdeviation of the main carrier frequency each time the 19 kHz pilotsub-carrier crosses the time axis. The second 38 kHz sub-carrier and itssidebands signal is the second 38 kHz sub-carrier double sideband,suppressed carrier, amplitude modulated by a four-channel, front minusback input signal, [(A+B)-(C+D)]. The second 38 kHz sub-carrier and itssidebands signal frequency modulates the main carrier 85% (excluding theSCA sub-carrier).

The 76 kHz sub-channel is a 76 kHz sub-carrier which is upper singlesideband, suppressed carrier, amplitude modulated by [(A+D)-(B+C)],which is a diagonal difference four-channel input signal component. The76 kHz sub-channel frequency modulates the main carrier 85×0.7=59.5%(excluding the SCA sub-carrier). The 76 kHz sub-channel consists of afirst and a second 76 kHz component. The first 76 kHz sub-channelcomponent results from modulating a first 76 kHz sub-carrier and thesecond component from modulating a second 76 kHz sub-carrier. The first76 kHz sub-carrier, sin2ω_(s) t, is the fourth harmonic of the 19 kHzpilot sub-carrier with the condition that each time the 19 kHz pilotsub-carrier crosses the time axis, the first 76 kHz sub-carrier crossesthe time axis simultaneously with a positive slope (increasing maincarrier frequency). The first 76 kHz sub-carrier is double sideband,suppressed carrier, amplitude modulated by the four-channel diagonaldifference input signal, [(A+D)-(B+C)], the result of which modulationis the first 76 kHz sub-carrier and its sidebands signal.

The second 76 kHz sub-carrier, cos2ω_(s) t, is the fourth harmonic ofthe 19 kHz pilot sub-carrier with the condition that each time the 19kHz sub-carrier crosses the time axis, the second 76 kHz sub-carriercauses an upward peak deviation of the main carrier. The second 76 kHzsub-carrier is double sideband, suppressed carrier, amplitude modulatedby the Hilbert transform of the four-channel diagonal difference inputsignal, namely, [(A+D)-(B+C)], the result of which modulation is thesecond 76 kHz sub-carrier and its sidebands signal.

The 76 kHz pilot sub-carrier, sin2ω_(s) t, is the fourth harmonic of the19 kHz pilot sub-carrier with the condition that each time the 19 kHzpilot sub-carrier crosses the time axis, the 76 kHz pilot sub-carriercrosses the time axis simultaneously with a positive slope (increasingmain carrier frequency). The 76 kHz pilot sub-carrier causes a 5%deviation of the main carrier.

The SCA component is a frequency modulated sub-carrier at a nominalcenter frequency of 67 kHz and modulates the main carrier 10%. When theSCA sub-channel is broadcast, the main channel, the first 38 kHzsub-carrier plus its sidebands signal, the second 38 kHz sub-carrierplus its sideband signal modulate the main carrier 75% while the 76 kHzsub-channel modulates the main carrier 75×0.7=52.5%.

The peak deviation of the main carrier resulting from simultaneousmodulation by the main channel, the first 38 sub-carrier and itssidebands signal, the second 38 kHz sub-carrier and its sidebandssignal, the 76 kHz sub-channel, the 19 kHz pilot sub-carrier, the SCAsub-channel, and the 76 kHz pilot sub-carrier is 100% of totalmodulation.

The pre-emphasis characteristics of all of the sub-carrier channels areidentical with those of the main channel (standard 75 microseconds).

The main channel and all sub-channels are capable of accepting audiofrequencies from 50 to 15,000

When only equal positive left-front and left-back signals are applied,the main channel modulation causes an upward deviation of the maincarrier frequency; also the first 38 kHz sub-carrier and its sidebandssignal crosses the time axis simultaneously with the first 38 kHzsub-carrier and in the same direction.

When only equal positive left-front and right-front signals are applied,the main channel modulation causes an upward deviation of the maincarrier frequency; also the second 38 kHz sub-carrier and its sidebandssignal crosses the time axis simultaneously with the 38 kHz sub-carrierand in the opposite direction.

When only equal positive left-front and right-back diagonal signals areapplied, the main channel modulation causes an upward deviation of themain carrier frequency; also the first 76 kHz sub-carrier and itssidebands signal crosses the time axis simultaneously with the first 76kHz sub-carrier in the opposite direction.

When only equal and increasing left-front and right-back diagonalsignals are applied, the main channel modulation causes an increasingmain carrier frequency; also the second 76 kHz sub-carrier and itssidebands signal crosses the time axis simultaneously with the second 76kHz sub-carrier and in the same direction.

The invention is not limited to the particular details of constructionof the embodiments depicted and other modifications and applications arecontemplated. For example, the described method and structure forminimizing the maximum amplitude of the baseband four-channel stereosignal when 38 kHz and 76 kHz signals are present simultaneously, andthus the maximum amplitude of the composite signal including the SCA andpilot signals under the same conditions, so as to minimize the maximumfrequency deviation of the RF carrier also under those same conditionshas been discussed in terms of its applicability to four-channel FMstereo communication, it has general applicability to any communicationsystem employing harmonically related sub-carriers which areamplitude-modulated, either double or single-sideband.

While particular embodiments of the present invention have been shownand described, it is apparent that changes and modifications may be madetherein without departing from the invention in its broader aspects. Theaim of the appended claims, therefore, is to cover all such changes andmodifications as fall within the true spirit and scope of the invention.

What is claimed is:
 1. A multi-channel stereo receiver for developing apredetermined plurality of discrete audio signals from a multi-channelcomposite stereo signal frequency modulated RF carrier, which compositesignal when initially created by a multiplexing system includes afour-element sum component representing the sum of four input audiosignals,a first two-element difference component representing adifference between elements of a related first pair of said audio inputsignals modulating a first subcarrier of angular frequency ω_(s), asecond two-element difference component representing a differencebetween elements of a related second pair of said audio input signalsmodulating a second subcarrier of angular frequency ω_(s) but displacedin phase, relative to the phase of said first subcarrier, by 90°, and afour-element difference component representing a difference between thesum of said first pair of audio input signals and the sum of said secondpair of audio input signals modulating a third subcarrier of angularfrequency ω_(s2), said receiver comprising: demodulating means forrecovering said composite signal from said modulated RF carrier; meansfor generating a like plurality of pulse trains; a demultiplexingsystem, having a like plurality of output terminals, coupled to saiddemodulating means and to said pulse generating means and responsive tosaid recovered composite signal and to said pulse trains for developinga like plurality of audio output signals individually issuing from anassigned one of said output terminals, each of said audio signalsincluding, in addition to a desired audio signal, undesired audioquantities; a like plurality of subtractors individually coupled toassigned ones of said demultiplexing system output terminals forindividually translating a selected one of said desired audio outputsignals to an assigned audio signal utilization circuit; and attenuatormeans coupled to the output of said demodulating means for selecting andapplying a portion of said recovered composite signal to each of saidsubtractors to adjust the levels of each of said desired audio signalstranslated through said subtractors to compensate for any change in thelevel of said desired audio signals attributable to said multiplexingsystem, as well as to substantially eliminate said undesired audioquantities from each of said desired audio signals.
 2. A stereo receiverof the type defined by claim 1 in which said portion of said recoveredcomposite signal applied to said subtractors by said attenuator is(1-(2/π) parts of said sum component.
 3. A multi-channel stereo receiverfor developing a predetermined plurality of discrete audio signals froma multi-channel composite stereo signal frequency modulated RF carrier,which composite signal when initially created by a multiplexing systemincludes a four-element sum component representing the sum of four inputaudio signals,a first two-element difference component representing adifference between elements of a related first pair of said audio inputsignals modulating a first subcarrier of angular frequency ω_(s), asecond two-element difference component representing a differencebetween elements of a related second pair of said audio input signalsmodulating a second subcarrier of angular frequency ω_(s) but displacedin phase, relative to the phase of said first sub-carrier, by 90°, and afour-element difference component representing a difference between thesum of said first pair of audio input signals and the sum of said secondpair of audio input signals modulating a third subcarrier of angularfrequency ω_(s2), said receiver comprising: demodulating means forrecovering said composite signal from said modulated RF carrier; meansresponsive to a timing signal for generating a switching signal; ademultiplexing system, having a like plurality of output terminals,coupled to said demodulating means and to said switching signalgenerating means and responsive to said recovered composite signal andto said switching signal for developing a like plurality of outputsignals individually issuing from an assigned one of said outputterminals, each of said output signals including, in addition to adesired audio difference signal, undesired audio quantities; a likeplurality of adders each having a first input terminal coupled to anassigned one of said demultiplexing system output terminals and eachfurther having a second input terminal and output terminal; first matrixmeans responsive to said recovered composite signal for deriving apreconditioning signal; means for coupling said preconditioning signalto said second input terminal of each of said adders to produce at theoutput of each of said adders modified audio difference signals; a likeplurality of similar phasing networks individually coupled to anassigned one of said adders' output terminals for effecting a firstpredetermined phase-vs-frequency characteristic of said modified audiodifference signals over a predetermined band of frequencies; meansresponsive to said recovered composite signal for deriving a quadratureoutput signal; a predetermined different phasing network responsive tosaid quadrature signal for effecting a predetermined differentphase-vs-frequency characteristic of said quadrature signal over apredetermined band of frequencies; and second matrix means for combiningsaid phase shifted modified audio difference signals and said phaseshifted quadrature signal to produce four discrete audio output signals.4. A receiver of the type defined by claim 3 in which said first matrixmeans derives a preconditioning signal of the form ((1/2π)-1/4)(A+B+C+D).
 5. A four-channel stereo receiver for developing a pluralityof two-element audio sum signals from a transmitted composite stereosignal frequency-modulating an RF carrier, which composite signaleffectively includes at least the following components,a four-elementsum component representing the sum of four input audio signals whichsignals are representative of first, second, third and fourth audiosources effectively located at the left-front, right-front, left-rearand right-rear of a listening point, a first two-element differencecomponent representing a difference between the elements of a diagonallyrelated first pair of said input audio signals modulating a firstsub-carrier of angular frequency ω_(s), a second two-element differencecomponent representing a difference between the elements of a diagonallyrelated second pair of said input audio signals modulating a secondsub-carrier of angular frequency ω_(s) but displaced in phase, relativeto the phase of said first sub-carrier, by 90°, and a pilot signalhaving an angular frequency ω_(s) /2 and a phase which is such that thephase of the second harmonic thereof is effectively displaced, relativeto the phase of said first sub-carrier, by 45°, said receivercomprising: demodulating means for recovering said composite signal fromsaid modulated RF carrier; decoding means responsive to said recoveredcomposite signal and to said pilot signal for deriving a plurality ofpredetermined audio difference components related to said two-elementdifference components of said composite stereo signal; a first matrixfor combining the four-element sum component of said recovered compositesignal with a first of said audio difference components to provide afirst audio sum component; a second matrix for combining said firstaudio sum component with a second of said audio difference components toprovide a first two-element audio sum signal; a third matrix forcombining the four-element sum component of said recovered compositesignal with a third of said audio difference components to provide asecond audio sum component; a fourth matrix for combining said secondaudio sum component with a fourth of said audio difference components toprovide a second two-element audio sum component; a fifth matrix forcombining said second audio sum signal with said second audio differencecomponent to provide a third two-element audio sum component; and meansfor utilizing said first, second and third two-element audio sumcomponents to simulate four channel sound reproduction.
 6. Afour-channel receiver of the type defined in claim 5 which furtherincludes means for modifying said third two-element audio sum componentsand a sixth matrix for combining the four element sum component of saidrecovered composite signal with said modified version of said thirdtwo-element audio sum component to provide a fourth two-element audiosum component.